Symbol-based signaling device for an electromagnetically-coupled bus system

ABSTRACT

The present invention provides an apparatus for transferring data through an electromagnetic coupler. The apparatus comprises a transmitter to encode a first plurality of bits into a symbol, a receiver to decode a transferred symbol into a second plurality of bits; and a coupling element having a geometry that provides robust electromagnetic transfer of the symbol and the transferred symbol. For one embodiment of the apparatus, the coupling element has a zig-zag geometry.

RELATED PATENT APPLICATIONS

This patent application is a continuation of U.S. patent applicationSer. No. 09/714,244, entitled “Symbol-Based Signaling For anElectromagnetically-Coupled Bus System” and filed on Nov. 15, 2000 nowU.S. Pat. No. 6,697,420.

BACKGROUND OF THE INVENTION

1. Technical Field

The present invention relates to mechanisms for communicating digitaldata, and in particular to mechanisms for communicating digital data inan electromagnetically-coupled bus system.

2. Background Art

Digital electronics systems, such as computers, must move data amongtheir component devices at increasing rates to take full advantage ofthe higher speeds at which these component devices operate. For example,a computer may include one or more processors that operate atfrequencies of a gigahertz (GHz) or more. The data throughput of theseprocessors outstrips the data delivery bandwidth of conventional systemsby significant margins. This discrepancy is mitigated somewhat byintelligent caching of data to maintain frequently used data on theprocessor chip. However, even the best caching architecture can leave aprocessor under-utilized. Similar problems arise in any digital system,such as communication networks, routers, backplanes, I/O buses, portabledevice interfaces, etc., in which data must be transferred among devicesthat operate at ever higher frequencies.

The digital bandwidth (BW) of a communication channel may be representedas:BW=F _(S)N_(S).Here, F_(S) is the frequency at which symbols are transmitted on achannel and N_(S) is the number of bits transmitted per symbol per clockcycle (“symbol density”). Channel refers to a basic unit ofcommunication, for example a board trace in single ended signaling orthe two complementary traces in differential signaling. For a typicalbus-based system, F_(S) is on the order of 200 MHz, N_(S) is one, andthe bus width (number of channels) is 32, which provides a bus data rateof less than one gigabyte per second.

Conventional strategies for improving BW have focused on increasing oneor both of the parameters F_(S) and N_(S). However, these parameterscannot be increased without limit. For example, a bus trace behaves likea transmission line for frequencies at which the signal wavelengthbecomes comparable to the bus dimensions. In this high frequency regime,the electrical properties of the bus must be carefully managed. This isparticularly true in standard multi-drop bus systems, which includethree or more devices that are electrically connected to each bus tracethrough parallel stubs. The connections can create discontinuities inthe trace impedance, which scatter high frequency signals. Interferencebetween scattered and unscattered signals can significantly reducesignal reliability. The resulting noise can be reduced through carefulimpedance matching of the system components. However, impedance matchingrequires the use of precision components, which increases the costs ofthese systems. In addition to impedance discontinuities, connections tobus traces may also affect system performance by adding capacitance.Capacitance can slow signal propagation speed and lower the traceimpedance, which may require larger driver circuits with increased powerconsumption.

Computer systems based on RAMBUS™ DRAM (RDRAM) technology representanother approach to high speed signaling. For these systems, devices aremounted on daughter cards, which are connected in series with the busthrough costly, tightly matched connectors. The impedance-matched seriesconnections eliminate the impedance discontinuities of parallel stubs,but the signal path must traverse each of the daughter cards, increasingits length. In addition, the different daughter card components must beimpedance matched to each other and the connectors, and the parasiticcapacitances of these components, all of which touch some portion of thebus, further affect the signal propagation speed, impedance, driversize, and power dissipation. These effects taken together seriouslyconstrain the total number of components (or bus capacity) that can beplaced on one bus.

Yet another strategy for addressing the frequency limits of conventionalbus systems is to replace the direct electrical connection between a bustrace and a device with an indirect, e.g. electromagnetic, coupling. Forexample, U.S. Pat. No. 5,638,402 discloses a system that employselectromagnetic couplers. The impact of an electromagnetic coupler onthe trace impedance depends strongly on the fraction of signal energy ittransfers between its coupling components, i.e. its couplingcoefficient. A coupler having a large coupling coefficient and/or lengthtransfers a large fraction of the signal energy it samples to itsassociated device. Large energy transfers can degrade the continuity ofthe trace impedance as much as standard direct electrical connections.They can also attenuate the signal energy rapidly, and on multi-dropbuses, little signal energy may be available to devices that are fartherfrom the signal source. On the other hand, coupling coefficients thatare too small or lengths that are too short result in low signal tonoise ratios at the devices. In addition, the coupling coefficient isvery sensitive to the relative positions of the coupling components.Variations in the relative positions can increase noise on the bus traceor reduce the transferred signal relative to non-scalable noiseaccording to whether the distance decreases or increases, respectively.

Practical BW limits are also created by interactions between the BWparameters, particularly at high frequencies. For example, the greaterself-induced noise associated with high frequency signaling limits thereliability with which signals can be resolved. This limits theopportunity for employing higher symbol densities.

Modulation techniques have been employed in some digital systems toencode multiple bits in each transmitted symbol, thereby increasingN_(S). Use of these techniques has been largely limited topoint-to-point communication systems, particularly at high signalingfrequencies. Because of their higher data densities, encoded symbols canbe reliably resolved only in relatively low noise environments.Transmission line effects limit the use of modulation in high frequencycommunications, especially in multi-drop environments. For example,RDRAM based systems may use four voltage levels (called QRSL) toincrease N_(S) to two. More aggressive modulation (amplitude modulationor other schemes) is precluded by the noise environment.

The present invention addresses these and other issues associated withcommunication of data in digital electronic systems.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention may be understood with reference to the followingdrawings, in which like elements are indicated by like numbers. Thesedrawings are provided to illustrate selected embodiments of the presentinvention and are not intended to limit the scope of the invention.

FIG. 1 is a block diagram of a conventional multi-drop bus system thatemploys electromagnetic couplers.

FIG. 2A is a block diagram of an electromagnetically-coupled multi-dropbus system in accordance with the present invention.

FIG. 2B is a block diagram representing the electrical properties of oneembodiment of the electromagnetically-coupled bus system of FIG. 2A.

FIGS. 3A-3E represent embodiments of the electromagnetic coupler ofFIGS. 2A and 2B, and their use in multi-drop bus systems.

FIG. 4 is a schematic representation of a symbol that representsmultiple bits of data through various modulation techniques that aresuitable for use with the present invention.

FIGS. 5A and 5B are block diagrams of embodiments of an interface thatis suitable for use with the present invention.

FIG. 6 is a block diagram of one embodiment of a transceiver module toencode and decode bits via amplitude, pulse-width, and phase modulation.

FIGS. 7A-7D are circuit diagrams for various components of oneembodiment of the transmitter of FIG. 6.

FIGS. 8A-8E represent signals at various stages of data transmission forone embodiment of bus system 200.

FIGS. 9A-9E are circuit diagrams for various components of oneembodiment of a receiver that is suitable for use with the presentinvention.

FIG. 10 is a block diagram representing a calibration circuit that issuitable for use with the present invention.

FIG. 11 is a frequency response plot of an embodiment of thecommunication channel of bus system 200.

DETAILED DESCRIPTION OF THE INVENTION

The following discussion sets forth numerous specific details to providea thorough understanding of the invention. However, those of ordinaryskill in the art, having the benefit of this disclosure, will appreciatethat the invention may be practiced without these specific details. Inaddition, various well-known methods, procedures, components, andcircuits have not been described in detail in order to focus attentionon the features of the present invention.

The present invention supports high bandwidth communication by providinggreater control over the frequency and encoding mechanisms employed totransfer data. A system in accordance with the present inventionincludes a data channel, such as a bus, having substantially uniformelectrical properties for transferring signals among devices that arecoupled through the data channel. The uniform electrical properties aresupported by an electromagnetic coupling scheme that allows higherfrequency signaling to be employed without significantly increasingnoise attributable to transmission line effects. The scheme employsbalanced electromagnetic couplers to provide reliable signal transferbetween the communication channel and the devices without significantlyimpacting the impedance of the communication channel. The resultingcleaner noise environment allows greater flexibility in selecting anencoding scheme to represent the data.

For one embodiment of the invention, a balanced electromagnetic couplerincludes first and second coupler components separated by a dielectricmedium and having a coupling coefficient in a specified range. At leastone of the coupler components has a geometry that reduces thesensitivity of the coupling coefficient to variations in the relativepositioning of the coupling components. The length of the coupler may beselected to provide sufficient signal energy transfer without limitingthe system bandwidth.

For another embodiment of the invention, devices transfer data to andfrom a multi-drop bus through electromagnetic couplers, using a selectedmodulation scheme. The electromagnetic couplers allow the devices tosample a relatively small portion of the signal energy on the bus, whichmitigates the impact of the devices on the electrical properties of thebus. The modulation schemes employed are selected to balance the symboldensity with susceptibility to inter and intra-symbol interference inthe impedance environment provided by the electromagnetically coupleddevices.

FIG. 1 is a block diagram representing the electrical properties of amulti-drop bus system 100. System 100 includes a bus 110 to transferdata among various devices 120(1)-120(n) (generically, “devices 120”).Device 120(1) is electrically coupled to bus 110, while devices120(2)-120(n) are coupled to bus 110 through associated electromagneticcouplers 160(1)-160(n−1), respectively. In the following discussion,electrical coupling refers to a relatively low resistance electricalpath between bus 110 and device 120(1) that is capable of transmittingsignals down to zero frequency (DC). Also shown in FIG. 1 are parasitics130, which may be associated with packages for devices 120 orconnectors, when devices 120 are provided on separate daughter cards.

For multi-drop bus systems, multiple electromagnetic couplers 160introduce impedance discontinuities along bus 110 that make impedancematching more difficult. Signals reflected from impedancediscontinuities interfere with other signals (inter-symbol andintra-symbol interference). The noise environment created by couplers160 and parasitics 130 (where present) limits the signaling frequenciesand the symbol densities that may be employed on system 100.

Electromagnetically coupled buses similar to system 100 are disclosed inU.S. Pat. Nos. 5,638,402, 3,516,065 and 3,619,504. The '402 patentdiscloses electromagnetic couplers 160 having parallel plate geometries(“parallel coupling portions”) and a “backward cross-talk coefficient”(K_(b)) of approximately 0.3. K_(b) represents the relative amplitude ofa counter-propagating signal induced across coupler 160 by a primarysignal. A K_(b) value of 0.3 implies strong signal scattering on bus 110and large signal energy loss per coupler. It also imposes large dynamicrange requirements on the receivers of devices 120. Even K_(b) values onthe order of 0.2 represent significant signal attenuations and noiseproblems on bus 110.

In addition to their strengths, the coupling coefficients of parallelplate couplers 160 are very sensitive to variations in horizontal (x, y)and vertical (z) alignment of the coupler components (162 and 164). Onesolution is to embed both sides of electromagnetic coupler 160 in acircuit board, with a precision sufficient to guarantee the couplingcoefficient falls in a targeted range. This precision increases thecosts of system 100. Moreover, it requires a connector, as representedby parasitics 130 to accommodate daughter cards.

Parallel plate couplers 160 are also susceptible to noise problems ifthey are implemented in a differential signaling scheme, wherecomplementary signals are driven on pairs of bus traces. For thesesystems, a pair of couplers 160 transfers the complementary signals to adifferential receiver in device 120. The sensitivity of parallel platecouplers 160 to variations in the positions of their componentsincreases the likelihood that coupler pairs have mismatched couplingcoefficients. This results in differential noise, which undermines thebenefits of differential signaling. Further, unless the couplers arespaced sufficiently far apart (increasing the circuit board area neededto support them), the complementary signals can cross couple, with aresulting loss in signal to noise ratio.

FIG. 2A is a block diagram representing one embodiment of a system 200in accordance with the present invention. System 200 may be a computersystem, but persons skilled in the art of digital communication andhaving the benefit of this disclosure will recognize that benefits ofthe present invention may be realized in any system that requires highbandwidth data transfers.

For system 200, devices 220(1)-220(m) (generically, “device 220”)communicate through a bus 210. For this purpose, devices 220(1)-220(m)include interfaces 230(1)-230( m), respectively, to transfer signals toand receive signals from bus 210. Interfaces 230(2)-230( m) communicatewith bus 210 through associated electromagnetic couplers240(1)-240(m−1), respectively (generically, “electromagnetic coupler240”). Electromagnetic couplers 240 are balanced to limit the impact ofdevices 220 on the electrical properties of bus 210, while providingreliable signal transmission between devices 220 and bus 210. Forexample, the coupling coefficients of electromagnetic couplers 240 areselected to transfer sufficient signal energy between bus 210 anddevices 220 to maintain signal to noise margins, while limiting signalreflections on bus 210 and the too rapid attenuation of signal energy onbus 210 (signal energy bleed-off). Balanced electromagnetic couplers 240typically employ coupling coefficients in the range of 0.1 to 0.4, e.g.K_(b)=0.05 to 0.2. The geometries of electromagnetic couplers 240 may bechosen to maintain these selected coupling coefficients againstvariations in the relative positioning of bus and device side couplingcomponents, 242 and 244, respectively (FIG. 2B).

Both the energy transferred by an electromagnetic coupler and themaximum effective signaling frequency supported by a system that employselectromagnetic couplers depend on the coupler length. In addition,longer couplers take up more space and entail larger system costs.

The signal energy transferred by coupler 240 is proportional theintegral of the square of the induced signal waveform over its duration.The induced signal waveform is determined by K_(b), the amplitude of thesignal waveform on the bus trace and the length of the coupler. For agiven value of K_(b), the longer the coupler, the more of the sampledsignal energy it transfers. In addition, if symbols are driven on bus210 at a sufficiently high frequency, the symbol period may be shorterthan the duration of the induced waveform. Under these circumstances,coupler 240 can mix energy from two or more symbols i.e. the symbolsinterfere, and this interference degrades the signal to noise ratio. Forthese reasons, the length of coupler 240 should be long enough toprovide adequate signal energy to the device without generatinginter-symbol interference or excessive energy bleed-off along bus 210.

FIG. 2B is a schematic representation of the electrical properties ofsystem 200. Signals are transmitted electromagnetically between adevice, e.g. device 220(2), and bus 210 through electromagnetic coupler240(1). In the following discussion, electromagnetic coupling refers tothe transfer of signal energy through the electric and magnetic fieldsassociated with the signal. Electromagnetic coupling includes both acapacitive component, associated with the electric field of the signal,and an inductive component, associated with the magnetic field of thesignal. For example, K_(b) is related to the inductive couplingcoefficient (K_(L)) and capacitive coupling coefficient (K_(C)) asfollows:K _(b)=0.25(K _(L) +K _(C))Here, K_(L) is the ratio of the mutual inductance per unit lengthbetween the coupler components to the geometric mean of theself-inductances of the coupler components, and K_(C) is the ratio ofthe mutual capacitance per unit length between the coupler components tothe geometric mean of the self-capacitances per unit length of thecoupler components.

The effects of the capacitive and inductive contributions on the energytransferred across coupler 240 vary with signal frequency. In general,the relative contribution of the inductive component becomes morepronounced with increasing signal frequency. For example, the presenceof a significant inductive component may be used to providedirectionality for signals at higher frequencies. In addition,electromagnetic coupler 240 behaves like a distributed device ratherthan a lumped device. The distributed nature of both capacitive andinductive aspects of coupler 240 become more pronounced at higherfrequencies, when the signal wavelengths become comparable to thephysical dimensions of coupler 240.

The use of electromagnetic couplers 240 with suitably selected couplingcoefficients significantly reduces the impedance discontinuities insystem 200 relative to those in systems that rely on electricalconnections or unbalanced electromagnetic couplers. Further, providingelectromagnetic couplers 240 with geometries that are relativelyinsensitive to variations in the positions of device and bus sidecomponents 242 and 244, respectively, allows the balanced couplingcoefficients to be maintained without need for costly, precisionmanufacturing. The more uniform impedance of bus 210 provides a cleanersignal environment in which to transmit data. Modulation schemesemployed to encode this data in accordance with the present inventionreflect both the cleaner noise environment of bus 210 and the effects ofcouplers 240 on the waveforms they transfer.

For one embodiment of the invention, electromagnetic coupler 240transfers approximately 5-10% of the signal amplitude on bus 210 to itscorresponding device 220. This corresponds to less than 1% of the signalenergy for a particular coupler geometry and length (K_(b)=0.13, L=1cm). The relatively small attenuation in signal energy on bus 210 limitsthe impact of multiple devices 220 on the impedance of bus 210. One sideeffect of this limited signal attenuation is that the signal waveform ondevice side 242 of electromagnetic coupler 240 (“transferred waveform”)is a small fraction of the energy transmitted on bus 210. Since thecoupling coefficient is symmetric, a similar attenuation occurs in thereverse direction, from device side 242 to bus 210. The significance ofthis signal attenuation depends on the types of noise in system 200.

Scalable noise is noise that scales with the energy of the signal.Scalable noise associated with the transferred waveform is attenuated tothe same extent as the transferred waveform itself. Sources of scalablenoise include signal reflections that are not eliminated byelectromagnetic coupler 240. Non-scalable noise includes externallycoupled noise, thermal noise, and the like. Signal attenuation byelectromagnetic coupler 240 may impact the performance of system 200 ifnon-scalable noise is not addressed. Strategies for addressingnon-scalable noise in system 200 include selecting robust symbolmodulation schemes and using differential signaling. For one embodimentof system 200, interface 230 amplifies the transferred waveform prior todemodulating it to recover the transmitted data.

Another side effect of electromagnetic coupler 240 is that thetransferred waveform is altered relative to the signal on bus 210. Ingeneral, a signal transferred across electromagnetic coupler 240 isdifferentiated. For example, a positive signal pulse 260 on bus side 244of electromagnetic coupler 240 becomes a positive/negative-going pulse270 on device side 242 of electromagnetic coupler 240. The modulationscheme(s) employed in system 200 is selected to accommodate theamplitude attenuation and signal differentiation associated withelectromagnetic couplers 240 without degrading the reliability of thecommunication channel. For example, signal attenuation, in the face ofnon-scaling noise sources, may limit the number of usable amplitudemodulation voltage levels. Differentiation may require the use ofintegration circuits to recover DC voltages for level signaling, if thatis desired instead of, or in addition to transition signaling. Also, theuse of rise-time modulation (described below) in system 200 entails themeasurement of the second derivative of a signal waveform.

For one embodiment of the invention, multi-drop bus system 200 is acomputer system and devices 220 correspond to various system components,such as processors, memory modules, system logic and the like. Anembodiment of the invention includes a 50 centimeter long bus 210 thatsupports up to 17 devices 220 capable of transferring data at a signalfrequency of 400 MHz. By employing modulation schemes that provide asymbol density of 4 bits per symbol, this embodiment of system 200provides a digital bandwidth of 1.6 gigabits per second per channel.Higher signal frequencies and higher symbol densities, enabled by therelatively clean noise environment of bus 210, may be employed toprovide even greater digital bandwidth. For example, using appropriatematerials, signaling frequencies on the order of 1 GHz may be employedin a multi-drop bus system.

FIG. 11 shows a family of curves that describe the bandwidth ofelectromagnetically coupled bus system 210 for the current state of theart in materials and electronic packages. The different curves representdifferent numbers of couplers and different coupling coefficients in atarget range. The shape is a bandpass filter with passband labeled 1101.The lower frequency bound is set by the frequency response of coupler240 and the upper bound is determined by printed circuit board materiallosses and package parasitic inductances and capacitances. Note that fora 1 cm. long coupler, the length-induced bandwidth limit occurs around 5GHz, but it occurs at lower frequencies for longer couplers, e.g. 1.25GHz for a 4 cm coupler length. Thus materials and parasitics limit theability to scale symbol frequency F_(S) higher. For example, theprevalent PC board dielectric material FR4 severely attenuatesfrequencies above 3 GHz. To increase digital bandwidth under theseconstraints, one is compelled to increase N_(S) by using modulationtechniques as described in the present invention. As materialcharacteristics are improved, for example by replacing FR4 with Teflon,the present invention can be scaled in F_(S), N_(S), or some combinationof the two to provide higher digital bandwidth as the passband 1101 ofbus system 210 is widened.

One advantage of the electromagnetic coupling between devices 220 andbus 210 is that devices 220 may be added to and removed from system 200more easily than in direct coupled systems or in electromagneticallycoupled systems that require precise positioning of the couplercomponents. For example, use of electromagnetic couplers 240 eliminatesthe need to make or break electrical connections to, for example, the 32traces of a 32-bit bus. Because of this, and benefits to electrostaticdischarge protection, signal integrity, etc, the electromagneticcoupling aspect of this invention may have important advantages toapplications such as hot-swapping.

For one embodiment of the present invention, electromagnetic couplers240 have geometries that make their coupling coefficients less sensitiveto the relative positioning of device side component 242 and bus sidecomponent 244. These geometries allow balanced couplers 240 to maintaintheir coupling coefficients in a selected range, despite variations inthe horizontal or vertical separations of device and bus side components242 and 244, respectively.

FIG. 3A represents one embodiment 300 of balanced electromagneticcoupler 240 having a geometry that provides relatively stable couplingbetween device 220 and bus 210. Coupler 300 is viewed looking in thenegative z direction, relative to the coordinate system indicated inFIG. 2B (a portion of which is reproduced in FIG. 3A). For thisorientation, a bus side component 320 appears above a device sidecomponent 330 of electromagnetic coupler 300. The geometries of bus anddevice side components 320, 330 allow the amount of energy transferredthrough coupler 300 to be relatively insensitive to the relativealignment of bus and device side components 320, 330.

For coupler 300, bus side component 320 undulates about a longitudinaldirection defined by its end-points (along the y-axis) to form a zig-zagpattern. The disclosed embodiment of bus side component 320 includesfour excursions from the longitudinal direction that alternate in thepositive and negative x direction. The disclosed number, size, andangles of the excursions from the longitudinal direction are provided toillustrate the geometry generally. Their values may be varied to meetthe constraints of a particular embodiment. Device side component 330has a similar zig-zag pattern that is complementary to that of bus sidecomponent 320.

The repeated crossings form parallel plate regions 340(1)-340(4)(generically, “parallel plate regions 340”) and fringe regions350(1)-350(3) (generically, “fringe regions 350”) for coupler 300.Parallel plate and fringe regions 340 and 350, respectively, providedifferent contributions to the coupling coefficient of coupler 300,which mitigate the effects of variations in the relative alignment ofcomponents 320 and 330. For example, the sizes of plate regions 340 donot vary significantly if components 320 and 330 are shifted slightlyfrom their reference positions in the x, y plane, and the sizes offringe regions 350 vary so that changes in adjacent regionsapproximately offset each other when components 320 and 330 are shiftedfrom their reference positions in the x, y plane. For an embodiment ofcoupler 300 in which S is 0.125 cm, δ=35°, and W is 5 mils, K_(C) variesby only ±2% as components 320 and 330 are shifted by ±8 mils in the xand/or y directions from their nominally aligned positions.

The effects of variations in the vertical separations between components320 and 330 are also mitigated in coupler 300. Coupling in parallelplate regions 340 varies inversely with separation (z), while variationsin fringe regions 350 vary more slowly with separation. The net effectis a reduced sensitivity to variations in z for coupler 300. With thischoice of coupler geometry, a +/−30% change in coupler separation (z)results in the capacitive coupling coefficient varying by less than+/−15%. This compares favorably with parallel plate based couplergeometries, which show a +40/−30% variation over the same range ofconductor separations.

For the disclosed embodiment of coupler 300, components 320 and 330 haverounded corners to provide a relatively uniform impedance environmentfor signals transmitted along either component. For similar reasons,components 320 and 330 have relatively uniform cross sections. In sum,coupler 300 provides robust signal transmission between device 220 andbus 210, without introducing significant impedance changes in eitherenvironment.

FIG. 3B represents another embodiment 304 of balanced electromagneticcoupler 240. For the disclosed embodiment, one component 324 retains theundulating or zig-zag geometry similar to that described above forcomponent 320, while a second component 334 has a substantially straightgeometry. Component 334 may form either the bus side or device side ofcoupler 304, while component 324 forms the opposite side. Coupler 304includes both parallel plate regions 344 and fringe regions 354,although the latter is smaller than fringe region 350 in coupler 300.Consequently, coupler 304 may be more sensitive to variations in therelative positions of components 324 and 334 than coupler 300.

FIG. 3C represents yet another embodiment 308 of balancedelectromagnetic coupler 240. For this embodiment, one component 328 isnarrower than a second component 338 to provide both parallel plateregion 348 and fringe regions 358.

FIG. 3D illustrates a portion of a multi-drop bus system 360 thatincorporates coupler 300. A bus trace 380 includes multiple bus sidecomponents 320 at spaced intervals along its length. Correspondingdevices 370 are coupled to bus trace 380 through their associated deviceside components 330. Components 320, 330 are shown rotated to indicatetheir geometry. Embodiments of coupler 300 may include selecteddielectric materials between components 320, 330 to facilitatepositioning or adjust the coupling coefficient.

FIG. 3E illustrates one mechanism for coupling device 370 to bus trace380. For the disclosed embodiment, bus trace 380, including bus sidecomponent 320 of coupler 300, is mounted on a circuit board 384. One endof bus trace 380 is connected to device 220(1). Device 370 is mounted ona flex circuit 386 and connected to device side component 330, only aportion of which is visible in FIG. 3E. Device side component 330continues along a surface of flex circuit 386 that faces bus sidecomponent 320 when flex circuit 386 is pressed against circuit board 384(as indicated by the arrow). A socket 388, only part of which is visiblein FIG. 3E, is provided to hold flex circuit 386 in place.

The flexible character of flex circuit 386 allows it to bend as it ispressed against circuit board 384. For one embodiment, device sidecomponent 330 resides on a relatively flat portion of flex circuit 386formed by pressing flex circuit 386 against circuit board 384. Whenfully inserted, looking down on coupler 300 along the negative z-axis,device side component 330 and bus side component 320 are aligned as inFIG. 3A. A spacer may be provided to maintain a separation between busand device side components 320 and 330, respectively, or one or both ofcomponents 320 and 330 may be coated with a dielectric material,allowing them to be pressed together without creating a short circuit. Atrace that couples device side component 330 to device 370 bends withflex circuit 386, eliminating the need for a connector between deviceside component 330 and device 370.

Flex circuit 386 may comprise, for example, one or more layers of aflexible and/or resilient material such as an epoxy dielectric material,a polyimide (e.g. Kapton® by E. I. du Pont de Nemours of Wilmington,Del.), or polyethylene terephthalate (PET). For one embodiment, deviceside component 330 may be sandwiched between layers of theflexible/resilient material, to provide the elasticity and dielectricisolation used to form coupler 300. The disclosed mechanism is just oneof many ways that may be used to couple device 370 to bus trace 380. Forexample, various combinations of flexible and rigid materials, daughtercards and variations on these mechanisms may be employed.

The cleaner noise environment provided by a multi-drop bus system thatis implemented in accordance with the present invention allows signalsto be transmitted reliably at higher frequencies than in conventionalmulti-drop bus systems. However, gains in bandwidth provided by highersignaling frequencies alone are limited. For example, the scale ofirregularities capable of scattering signals in the transmission channeldecreases as the signal frequency increases. In addition, parasiticcapacitances and inductances, which can not be completely eliminated ormasked, reduce transmission speed, attenuate signal amplitudes, andcreate circuit resonances at high frequencies. Further, materialproperties such as skin effect and dielectric losses may limit highfrequency propagation. The signal attenuation by electromagnetic coupler240 may also affect bandwidth. For example, amplifying transmittedsignals to offset attenuation may limit the frequency at which signalscan be transmitted.

As noted above, the digital bandwidth of a channel is given by BW=F_(S)N_(S), where F_(S) is the symbol frequency and N_(S) is the number ofbits transmitted per symbol (“symbol density”). For one embodiment ofthe present invention, various modulation schemes are employed toincrease N_(S), for a specified symbol period (1/F_(S)). For a givenF_(S), the larger N_(S) provides an overall increase in BW that avoidsthe limitations associated with reliance on high frequency signalingalone. Selected modulation schemes may be combined with high frequencysignaling to provide significant increases in BW.

In the following discussion, various time-domain modulation schemes areused for purposes of illustration. The benefits of the present inventionare not limited to the disclosed modulation schemes. Other time-domainmodulation schemes, such as shape modulation (varying the number ofedges in a pulse), narrowband and wideband frequency-domain modulationschemes, such as frequency modulation, phase modulation, and spreadspectrum, or combinations of both time and frequency-domain modulationschemes (a pulse superposed with a high frequency sinusoid), are alsosuitable for use with this invention.

FIG. 4 is a schematic representation of a signal 410 that illustratesthe interplay between F_(S), N_(S), and various modulation schemes thatmay be employed to encode multiple data bits into a symbol. Signal 410includes a modulated symbol 420 transmitted in a symbol period (F_(S)⁻¹). For purposes of illustration, phase, pulse-width, rise-time, andamplitude modulation schemes are shown encoding five bits of data(N_(S)=5) in symbol 420. The present invention may implement thesemodulation schemes as well as others, alone or in combination, toincrease the bandwidth for a particular system. The modulation scheme(s)may be selected by considering the bit interval (see below), noisesources, and circuit limitations applicable to each modulation schemeunder consideration, and the symbol period available for a givenfrequency.

In the following discussion, a “pulse” refers to a signal waveformhaving both a rising edge and a falling edge. For pulse-based signaling,information may be encoded, for example, in the edge positions, edgeshapes (slopes), and signal amplitudes between edge pairs. The presentinvention is not limited to pulse-based signaling, however. Other signalwaveforms, such as edge-based signaling and various types of amplitude,phase, or frequency-modulated periodic waveforms may be implemented aswell. The following discussion focuses on modulation of pulse-basedsignaling schemes to elucidate various aspects of the present invention,but these schemes are not necessary to practice the invention.Considerations similar to those discussed below for pulse-basedsignaling may be applied to other signal waveforms to select anappropriate modulation scheme.

For signal 410, the value of a first bit (0 or 1) is indicated by where(p₀ or p₁) the leading edge of symbol 420 occurs in the symbol period(phase modulation or PM). The values of second and third bits areindicated by which of 4 possible widths (w₀, w₁, w₂, w₃) the pulse has(pulse-width modulation or PWM). The value of a fourth bit is indicatedby whether the falling edge has a large (rt₀) or small (rt₁) slope(rise-time modulation or RTM), and the value of a fifth bit is indicatedby whether the pulse amplitude is positive or negative (a₀, a₁)(amplitude modulation or AM). Bold lines indicate an actual state ofsymbol 420, and dashed lines indicate other available states for thedescribed encoding schemes. A strobe is indicated within the symbolperiod to provide a reference time with which the positions of therising and falling edges may be compared. The number of bits encoded byeach of the above-described modulation schemes is provided solely forillustration. In addition, RTM may be applied to the rising and/or isfalling edges of symbol 420, and AM may encode bits in the magnitudeand/or sign of symbol 420.

PM, PWM, and RTM are examples of time-domain modulation schemes. Eachtime-domain modulation scheme encodes one or more bits in the time(s) atwhich one or more events, such as a rising edge or a rising edgefollowed by a falling edge, occur in the symbol period. That is,different bit states are represented by different event times ordifferences between event times in the symbol period. A bit intervalassociated with each time-domain modulation scheme represents a minimumamount of time necessary to reliably distinguish between the differentbit states of the scheme. The modulation schemes selected for aparticular system, and the number of bits represented by a selectedmodulation scheme is determined, in part, by the bit intervals of thecandidate modulation schemes and the time available to accommodate them,i.e. the symbol period.

In FIG. 4, t₁ represents a minimum time required to distinguish betweenp₀ and p₁ for a phase modulation scheme. One bit interval of duration t₁is allocated within the symbol period to allow the pulse edge to bereliably assigned to p₀ or p₁. The value of t₁ depends on noise andcircuit limitations that can interfere with phase measurements. Forexample, if the strobe is provided by a clock pulse, clock jitter maymake the strobe position (time) uncertain, which increases the minimuminterval necessary to reliably distinguish between p₀ and p₁. Variouscircuit limitations and solutions are discussed below in greater detail.

Similarly, one bit interval of duration t₃ is allocated within thesymbol period to allow the two states (rt₀, rt₁) to be distinguishedreliably. The size of t₃ is determined by noise and circuit limitationsassociated with rise time measurements. For example, rise times aredifferentiated by passing through coupler 240. Consequently, t₃ must belong enough to allow the measurement of a second derivative.

Three bit intervals of duration t₂ are allocated within the symbolperiod to allow the four states (w₀, w₁, w₂, W₃) to be reliablydistinguished. The size of t₂ is determined by noise and circuitlimitations associated with pulse width measurements. If pulse width isdetermined relative to a clock strobe, considerations regarding clockjitter may apply. If pulse width is determined relative to, e.g., theleading edge of a pulse, considerations such as supply voltagevariations between the measurements of the leading and trailing edgesmay apply.

In general, the time needed to encode an n-bit value in a time-domainmodulation scheme (i) that has a bit interval, t_(i), is(2^(n)−1)·t_(i). If non-uniform bit intervals are preferred for noise orcircuit reasons, the total time allotted to a modulation scheme is thesum of all of its bit intervals. When multiple time-domain modulationschemes are employed, the symbol period should be long enough toaccommodate Σ (2^(n(i))−1)·t_(i), plus any additional timing margins.Here, the summation is over all time-domain modulation schemes used. Inthe above example, the symbol period should accommodate t₁+t₃+3 t₂, plusany other margins or timings. These may include minimum pulse widthsindicated by channel bandwidth, residual noise, and the like.

Using multiple encoding schemes reduces the constraints on the symboltime. For example, encoding 5 bits using pulse width modulation alonerequires at least 31·t₂. If t₂ is large enough, the use of the singleencoding scheme might require a larger symbol period (lower symbolfrequency) than would otherwise be necessary.

A minimum resolution time can also be associated with amplitudemodulation. Unlike the time domain modulation schemes, amplitudemodulation encodes data in pulse properties that are substantiallyorthogonal to edge positions. Consequently, it need not add directly tothe total bit intervals accommodated by the symbol period. For example,amplitude modulation uses the sign or magnitude of a voltage level toencode data.

The different modulation schemes are not completely orthogonal, however.In the above example, two amplitude states (positive and negative)encode one bit, and the minimum time associated with this interval maybe determined, for example, by the response time of a detector circuitto a voltage having amplitude, A. The pulse width should be at leastlong enough for the sign of A to be determined. Similarly, a symbolcharacterized by rise-time state rt₁ and width state w₃ may interferewith a next symbol characterized by phase state p₀. Thus, noise andcircuit limitations (partly summarized in the bit intervals), therelative independence of modulation schemes, and various other factorsare considered when selecting modulation schemes to be used with thepresent invention.

FIG. 5A is a block diagram of an embodiment 500 of interface 230suitable for processing multi-bit symbols for devices 220(2)-220(m). Forexample, interface 500 may be used to encode outbound bits from, e.g.,device 220(2) into a corresponding symbol for transmission on bus 210,and to decode a symbol received on bus 210 into inbound bits for use bydevice 220(2).

The disclosed embodiment of interface 230 includes a transceiver 510 anda calibration circuit 520. Also shown in FIG. 5A is device sidecomponent 242 of electromagnetic coupler 240 to provide a transferredwaveform to transceiver 510. For example, the transferred waveform maybe the differentiated waveform generated by transmitting pulse 420across electromagnetic coupler 240. A device side component 242 isprovided for each channel, e.g. bus trace, on which interface 230communicates. A second device side component 242′ is indicated for thecase in which differential signaling is employed.

Transceiver 510 includes a receiver 530 and a transmitter 540. Receiver530 recovers the bits encoded in the transferred waveform on device sidecomponent 242 of electromagnetic coupler 240 and provides the recoveredbits to the device associated with interface 230. Embodiments ofreceiver 530 may include an amplifier to offset the attenuation ofsignal energy on transmission across electromagnetic coupler 240.Transmitter 540 encodes data bits provided by the associated device intoa symbol and drives the symbol onto device side 242 of electromagneticcoupler 240.

Calibration circuit 520 manages various parameters that may impact theperformance of transceiver 510. For one embodiment of interface 230,calibration circuit 520 may be used to adjust termination resistances,amplifier gains, or signal delays in transceiver 510, responsive tovariations in process, temperature, voltage, and the like.

FIG. 5B is a block diagram of an embodiment 504 of interface 230 that issuitable for processing encoded symbols for a device that is directlyconnected to the communication channel. For example, in system 200 (FIG.2), device 220(1) may represent the system logic or chipset of acomputer system that is directly connected to a memory bus (210), anddevices 220(2)-220(m) may represent memory modules for the computersystem. Accordingly, a DC connection 506 is provided for each channel ortrace on which interface 504 communicates. A second DC connection 506′(per channel) is indicated for the case in which differential signalingis employed. Interface 504 may include a clock synchronization circuit560 to account for timing differences in signals forwarded fromdifferent devices 220(2)-220(m) and a local clock.

FIG. 6 is a block level diagram representing an embodiment 600 oftransceiver 510 that is suitable for handling waveforms in which databits are encoded using phase, pulse-width and amplitude modulation, andthe strobe is provided by a clock signal. Transceiver 600 supportsdifferential signaling, as indicated by data pads 602, 604, and itreceives calibration control signals from, e.g., calibration circuit520, via control signals 608.

For the disclosed embodiment of transceiver 510, transmitter 540includes a phase modulator 640, a pulse-width modulator 630, anamplitude modulator 620 and an output buffer 610. Output buffer 610provides inverted and non-inverted outputs to pads 602 and 604,respectively, to support differential signaling. A clock signal isprovided to phase modulator 640 to synchronize transceiver 510 with asystem clock. The disclosed configuration of modulators 620, 630, and640 is provided only for purposes of illustration. The correspondingmodulation schemes may be applied in a different order or two or moreschemes may be applied in parallel.

The disclosed embodiment of receiver 530 includes an amplifier 650, anamplitude demodulator 660, a phase demodulator 670, and a pulse-widthdemodulator 680. The order of demodulators 660, 670, and 680 is providedfor illustration and is not required to implement the present invention.For example, various demodulators may operate on a signal in parallel orin an order different from that indicated.

Devices 690(a) and 690(b) (generically, “device 690”) act as on-chiptermination impedances, which in one embodiment of this invention areactive while interface 230 is receiving. The effectiveness of device 690in the face of, e.g., process, temperature, and voltage variations maybe aided by calibration circuit 520. For transceiver 600, device 690 isshown as an N device, but the desired functionality may be provided bymultiple N and/or P devices in series or in parallel. The controlprovided by calibration circuit 520 may be in digital or analog form,and may be conditioned with an output enable.

FIG. 7A is a circuit diagram of one embodiment of transmitter 540 andits component modulators 620, 630, 640. Also shown is a strobetransmitter 790 suitable for generating a strobe signal, which may betransmitted via bus 210. For one embodiment of system 200, two separatestrobes are provided. One strobe is provided for communications fromdevice 220(1) to devices 220(2) through 220(m), and another strobe isprovided for communications from devices 220(2) through 220(m) back todevice 220(1).

The disclosed embodiment of transmitter 540 modulates a clock signal(CLK_PULSE) to encode four outbound bits per symbol period. One bit isencoded in the symbol's phase (phase bit), two bits are encoded in thesymbol's width (width bits) and one bit is encoded in the symbol'samplitude (amplitude bit). Transmitter 540 may be used to generate adifferential symbol pulse per symbol period, and strobe transmitter 790may be used to generate a differential clock pulse per symbol period.

Phase modulator 640 includes a MUX 710 and delay module (DM) 712. MUX710 receives a delayed version of CLK_PULSE via DM 712 and an undelayedversion of CLK_PULSE from input 704. The control input of MUX 710transmits a delayed or undelayed first edge of CLK_PULSE responsive tothe value of the phase bit. In general, a phase modulator 640 thatencodes p-phase bits may select one of 2^(P) versions of CLK_PULSEsubject to different delays. For the disclosed embodiment, the output ofphase modulator 640 indicates the leading edge of symbol 420 and servesas a timing reference for generation of the trailing edge by widthmodulator 630. A delay-matching block (DMB) 714 is provided to offsetcircuit delays in width modulator 630 (such as the delay of MUX 720)which might detrimentally impact the width of symbol 420. The output ofDMB 714 is a start signal (START), which is provided to amplitudemodulator 620 for additional processing.

Width modulator 630 includes DMs 722, 724, 726, 728, and MUX 720 togenerate a second edge that is delayed relative to the first edge by anamount indicated by the width bits. The delayed second edge forms a stopsignal (_STOP) that is input to amplitude modulator 620 for additionalprocessing. For the disclosed embodiment of transmitter 540, two bitsapplied to the control input of MUX 720 select one of four differentdelays for the second edge, which is provided at the output of MUX 720.Inputs a, b, c, and d of MUX 720 sample the input signal, i.e. the firstedge, following its passage through DMs 722, 724, 726, and 728,respectively. If the width bits indicate input c, for example, thesecond edge output by MUX 720 is delayed by DM 722+DM 724+DM 726relative to the first edge.

Amplitude modulator 620 uses START and _STOP to generate a symbol pulsehaving a first edge, a width, and a polarity indicated by the phase,width, and amplitude bits, respectively, provided to transmitter 540 fora given symbol period. Amplitude modulator 620 includes switches 740(a)and 740(b) which route START to edge-to-pulse generators (EPG) 730(a)and 730(b), respectively, depending on the state of the amplitude bit.Switches 740 may be AND gates, for example. _STOP is provided to secondinputs of EPGs 730(a) and 730(b) (generically, EPG 730). On receipt ofSTART, EPG 730 initiates a symbol pulse, which it terminates on receiptof _STOP. Depending on which EPG 730 is activated, a positive or anegative going pulse is provided to the output of transmitter 540 viadifferential output buffer 610.

Strobe transmitter 790 includes DM 750 and matching logic block 780. DM750 delays CLK_PULSE to provide a strobe signal that is suitable forresolving the data phase choices p0 and p1 of symbol 420. For oneembodiment of strobe transmitter 790, DM 750 positions the strobe evenlybetween the phase bit states represented by p0 and p1 (FIG. 4). Thestrobe is used by, e.g., receiver 530 to demodulate phase by determiningif the leading edge of data arrives before or after the strobe. DM 750of strobe transmitter 790 thus corresponds to phase modulator 640 ofdata transmitter 540. Matching logic block 780 duplicates the remainingcircuits of transmitter 540 to keep the timing of the strobe consistentwith the data, after DM 750 has fixed the relative positioning.

In general, DM 750 and matching logic block 780 duplicate for the strobethe operations of transmitter 540 on data signals at the level ofphysical layout. Consequently, this delay matching is robust tovariations in process, temperature, voltage, etc. In addition, theremainder of the communication channel from the output of transmitter540, through board traces, electromagnetic coupler 240, board traces onthe other side of coupler 240, and to the inputs of receiver 530 at thereceiving device, may be matched in delays between data and strobe inorder to keep the chosen relative timing. However, the matching ofdelays is one embodiment described for illustrative purposes and is notnecessary to practice this invention. For example, if the circuits andremainder of the channel do not maintain matched data to strobe delays,receivers may calibrate for the relative timing of the strobe or evencompensate for the absence of a strobe by recovering the timing fromappropriately encoded data.

FIG. 7B is a schematic diagram of one embodiment of a programmable delaymodule (DM) 770 that is suitable for use with the present invention. Forexample, one or more DMs 770 may be used for any of DMs 712, 722, 724,726, 728, and 750 in the disclosed embodiment of transmitter 540 tointroduce programmable delays in START and _STOP. DM 770 includesinverters 772(a) and 772(b) that are coupled to reference voltages V₁and V₂ through first and second transistor sets 774(a), 774(b) and776(a), 776(b), respectively. Reference voltages V₁ and V₂ may be thedigital supply voltages in some embodiments. Programming signals,p_(i)-p_(j) and n₁-n_(k), applied to transistor sets 774(a), 774(b) and776(a), 776(b), respectively, alter the conductances seen by inverters772(a) and 722(b) and, consequently, their speeds. As discussed below ingreater detail, calibration circuit 520 may be used to selectprogramming signals, p₁-p_(j) and n₁-n_(k), for inverters 772(a) and772(b).

FIG. 7C is a schematic diagram of one embodiment of EPG 730 that issuitable for use with the present invention. The disclosed embodiment ofEPG 730 includes transistors 732, 734, and 736 and inverter 738. Thegate of N-type transistor 734 is driven by START. A positive-going edgeon START indicates the beginning of a symbol pulse. The gates of P andN-type transistors 732 and 736, respectively, are driven by _STOP,which, for EPG 730(a) and 730(b) in FIG. 7A, is a delayed, inverted copyof START. A negative-going edge on _STOP indicates the end of a symbolpulse. When_STOP is high, transistor 732 is off and transistor 736 ison. A positive-going edge on START turns on transistor 734, pulling nodeN low and generating a leading edge for a symbol pulse at the output ofEPG 730. A subsequent negative-going edge on _STOP, turns off transistor736 and turns on transistor 732, pulling node N high and terminating thesymbol pulse.

For a given symbol pulse, START may be deasserted (negative-going edge)before or after the corresponding _STOP is asserted. For example, thedisclosed embodiment of transmitter 540 is timed with CLK_PULSE, andhigher symbol densities may be obtained by employing narrow CLK_PULSEs.The widths of START and _STOP are thus a function of the CLK_PULSEwidth, while the separation between START and_STOP is a function of thewidth bits. The different possible relative arrivals of the end of STARTand beginning of _STOP may adversely impact the modulation of symbol 420by the width bits. Specifically, transistor 734 may be on or off when anegative-going edge of _STOP terminates the symbol pulse. Node N maythus either be exposed to the parasitic capacitances at node P throughtransistor 734, or not. This variability may affect the delay of thetrailing symbol edge through EPG 730 in an unintended way.

FIG. 7D is a schematic diagram of an alternative embodiment oftransmitter 540 that includes an additional EPG 730(c). EPG 730(c)reshapes START to ensure a consistent timing which avoids thevariability described above. Namely, the modified START is widened sothat it always ends after _STOP begins. This is done by generating a newSTART whose beginning is indicated by the original START but whose endis indicated by the beginning of _STOP, instead of the width ofCLK_PULSE. Note also that, in the alternative embodiment shown in FIG.7D, the sum of the delays through delay matching block 714 and EPG730(c) must match the unintended delays in width modulator 630.

FIG. 8A-8E show CLK_PULSE, START, STOP, SYMBOL, and TR_SYMBOL,respectively, for one embodiment of system 200. Here, TR_SYMBOLrepresents the form of SYMBOL following transmission acrosselectromagnetic coupler 240. The smaller amplitude of TR_SYMBOL relativeto SYMBOL is roughly indicated by the scale change between the waveformsof FIG. 8D and 8E. TR_SYMBOL represents the signal that is decoded byinterface 230 to extract data bits for further processing by device 220.The 4 outbound bits encoded by each SYMBOL are indicated below thecorresponding SYMBOL in the order (p, w₁, w₂, a).

FIG. 9A is a schematic diagram representing one embodiment of receiver530 that is suitable for use with the present invention. The disclosedembodiment of receiver 530 processes differential data signals. FIG. 9Aalso shows a strobe receiver 902, which is suitable for processing adifferential strobe signal. Strobe receiver 902 may provide delaymatching for receiver 530 similar to that discussed above. Receiver 530and strobe receiver 902 may be used, for example, in system 200 inconjunction with the embodiments of transmitter 540 and strobetransmitter 790 discussed above.

The disclosed embodiment of receiver 530 includes differential tosingle-ended amplifiers 920(a) and 920(b) which compensate for theenergy attenuation associated with electromagnetic coupler 240.Amplifiers 920(a) and 920(b) produce digital pulses in response toeither positive or negative pulses on the transferred signal (TR_SYMBOLin FIG. 8E) and its complement, e.g. the signals at inputs 602 and 604.In addition to amplification, amplifiers 920 may latch their outputswith appropriate timing signals to provide sufficient pulse widths forsucceeding digital circuits.

Matching strobe receiver 902 similarly amplifies the accompanyingdifferential strobe signal. For the disclosed embodiment, the receivedstrobe is used to decode phase information in data symbol 420. Strobereceiver 902 includes differential to single-ended amplifiers 920(c) and920(d) and matched circuitry 904. Matched circuitry 904 replicates muchof the remaining circuitry in receiver 530 to match delays for data andstrobe signals, similar to the matching of transmitter 540 and strobetransmitter 790. One embodiment of strobe receiver 902 includes circuitsthat correspond to phase demodulator 670 and width demodulator 680 withsome minor modifications. For example, strobe buffer 990 buffers thereceived strobe for distribution to multiple receivers 530, up to thenumber of channels in, e.g., bus 210. Strobe buffer 990 may be large,depending on the number of receivers it drives. Data buffer 980corresponds to strobe buffer 990. To save area, data buffer 980 need notbe an exact replica of strobe buffer 990. The delays can also be matchedby scaling down both data buffer 980 and its loading proportionately,relative to their counterparts in strobe receiver 902.

Uni-OR gate (UOR) 940(a) combines the outputs of amplifiers 920(a) and920(b) to recover the first edge of TR_SYMBOL. The name uni-OR indicatesthat the propagation delay through gate 940 is uniform with respect tothe two inputs. An embodiment of UOR 940 is shown in FIG. 9C. Similarly,uni-AND gate (UAND) 930 recovers the second edge of TR_SYMBOL. Anembodiment of UAND 930 is shown in FIG. 9B.

The disclosed embodiment of phase demodulator 670 includes an arbiter950(b) (generically, “arbiter 950”) and data buffer 980. Arbiter 950(b)compares the first edge recovered from the transferred symbol by UOR940(a) with the corresponding edge from the recovered strobe by UOR940(b), respectively, and sets a phase bit according to whether therecovered first edge of the symbol leads or follows the first edge ofthe strobe. An embodiment of arbiter 950 is shown in FIG. 9D. An output952 goes high if input 956 goes high before input 958. Output 954 goeshigh if input 958 goes high before input 956.

FIG. 9E is a circuit diagram representing one embodiment of amplifier920. The disclosed embodiment of amplifier 920 includes a resetequalization device 922, a gain control device 924, and a pre-chargedlatch 928. Reset device 922 speeds up the resetting of amplifier 920after a detected pulse, in preparation for the next symbol period. Gaincontrol device 924 compensates the gain of amplifier 920 for variationsin process, voltage, temperature, and the like. A control signal 926 maybe provided by calibration circuit 520. More generally, device 924 maybe multiple devices connected in series or parallel, and signal 926 maybe several bits produced by calibration circuit 520. Pre-charged latch928 reshapes received pulses for the convenience of succeeding circuits.Resulting output pulse widths are determined by a timing signal, _RST.For one embodiment of amplifier 920, _RST is produced by DM 916 (FIG.9A), along with other timing signals used in receiver 530. It ispossible for pre-charged latch 928 and signal _RST to be in inconsistentstates, due to power-on sequences or noise. Additional circuitry may beused to detect and correct such events.

The disclosed embodiment of amplitude demodulator 660 includes anarbiter 950(a) which receives the amplified transferred signals fromamplifiers 920(a) and 920(b). Arbiter 950(a) sets an amplitude bitaccording to whether the output of amplifier 920(a) or 920(b) pulsesfirst.

The disclosed embodiment of width demodulator 680 includes delay modules(DMs) 910, 912, 914, arbiters 950(c), 950(d), 950(e), and decoding logic960. The recovered first symbol edge is sent through DMs 910, 912, and914 to generate a series of edge signals having delays that replicatethe delays associated with different symbol widths. For one embodimentof the invention, DMs 910, 912, and 914 may be implemented asprogrammable delay modules (FIG. 7B). Arbiters 950(c), 950(d), and950(e) determine the (temporal) position of the second edge with respectto the generated edge signals. Decoding logic 960 maps this position toa pair of width bits.

Latches 970(a), 970(b), 970(c), and 970(d) receive first and secondwidth bits, the phase bit, and the amplitude bit, respectively, at theirinputs, and transfer the extracted (inbound) bits to their outputs whenclocked by a clocking signal. For the disclosed embodiment of receiver530, the latches are clocked by sampling a signal from the delay chainof width demodulator 680 through the extra delay of DM 916. Thislatching synchronizes the demodulated bits to the accompanying strobetiming. In addition, a device 220 may require a further synchronizationof the data to a local clock, e.g. clock synchronization circuit 560 inFIG. 5B. Persons skilled in the art and having the benefit of thisdisclosure will appreciate that this can be done in any number ofdifferent ways.

The various components in an embodiment of interface 230 include anumber of circuit elements that may be adjusted to compensate forprocess, voltage, temperature variations and the like. For example,compensation may entail adjusting the delay provided by a programmabledelay module (DM 770), the gain provided by an amplifier (amplifier920), or the termination resistance (device sets 690(a) and 690(b)).

FIG. 10 shows an embodiment of calibration circuit 520. The purpose ofcalibration is to use feedback to measure and compensate for variableprocess, temperature, voltage, and the like. The embodiment ofcalibration circuit 520 shown in FIG. 10 is a delay-locked loop (DLL). Aclock signal (CLK_PULSE) is delayed by series-connected DMs1000(1)-1000(m). The number of DMs is chosen so that the sum of thedelays can be set to match one period of CLK_PULSE. Arbiter 950 is usedto detect when the sum of the delays through DMs 1000 is less than,equal to, or more than one clock period. DLL control 1010 cycles throughdelay control settings until the sum of the delays matches one clockperiod. The established control setting reflects the effects of process,temperature, voltage, etc . . . on the delays of DMs 1000. Calibrationcircuit 520 may be operated continuously, periodically, when conditions(temperature, voltage, etc. ) change, or according to any of a varietyof other strategies.

The same calibration control settings can be distributed to DMs usedthroughout interface 230, such as DM 712, DM 910, etc. The desireddelays of DMs in interface 230 are achieved by selecting a number ofprogrammable delay modules 770 for each such DM which have the sameratio to the total number of delay modules 770 included in all the DMs1000 as the ratio of the desired delay to the clock period. For example,if there are 20 total delay modules 770 in the sum of DMs 1000, one canselect a delay of one tenth of the clock period by using 2 delay modules770 for any particular DM used in interface 230. In addition, one canalso choose a fractional extra delay for any particular DM by insertingsmall extra loads at the outputs of selected delay modules 770 whichconstitute that DM.

The calibration information obtained by calibration circuit 520 may alsobe used to control other circuit parameters, in the face of variableconditions. These may include the resistance of termination device 690and gain of amplifier 920. This may be done by correlating theinformation contained in the delay control setting with the effects oflike process, temperature, voltage, and like conditions on the othercircuit parameters.

There has thus been disclosed a mechanism for providing high bandwidthcommunications in multi-drop bus systems. The disclosed system employselectromagnetic couplers to transfer data to and from a multi-drop bus.The electromagnetic couplers impose relatively minor perturbations onthe electrical properties of the bus, reducing the noise associated withhigh frequency, transmission line effects. The cleaner noise environmentallows various modulation schemes to be implemented in multi-drop bussystems at higher signaling frequencies.

The disclosed embodiments have been provided to illustrate variousfeatures of the present invention. Persons skilled in the art ofbus-based system design, having the benefit of this disclosure, willrecognize variations and modifications of the disclosed embodiments,which none the less fall within the spirit and scope of the appendedclaims.

1. An apparatus comprising: a transmitter to encode plural bits intoselected properties of a symbol; a trace including an electromagneticcoupling element having a shape, size, and distance from a bus to enablea substantially reliable signal transfer between the electromagneticcoupling element and the bus without significantly changing theimpedance of the bus; a receiver to detect a transferred symbol inducedon the electromagnetic coupling element and to decode plural bitsrepresented by the transferred symbol; and a termination device that isresponsive to a calibration signal received by the apparatus.
 2. Anapparatus comprising: a flexible material; an electronic device; and aconductor mounted on the flexible material and coupled to. theelectronic device, the conductor including a coupling element having ashape, size, and distance from a bus to transfer symbolselectromagnetically to and from the electronic device withoutsignificantly changing the impedance of the bus.
 3. The apparatus ofclaim 2, wherein the electronic device comprises: a memory structure tostore digital data; and an interface to translate between the digitaldata and the symbols.
 4. The apparatus of claim 3, wherein the digitaldata includes plural bits and the interface comprises a transmitter totranslate the plural bits into a symbol for transmission through thecoupling element.
 5. The apparatus of claim 4, wherein the flexiblematerial is flexed to provide a curved portion and the coupling elementinterface is positioned on the curved portion.
 6. The apparatus of claim5, wherein the curved portion of the flexible material acquires asubstantially planar geometry at the position of the coupling element,if the apparatus is pressed against a substantially planar surface. 7.The apparatus of claim 6, further comprising a mechanism to hold theapparatus against the substantially planar surface.
 8. The apparatus ofclaim 2, wherein the coupling element has a zig-zag geometry.
 9. Amemory module comprising: a memory device to store a plurality of bits;a transmitter to encode the plurality of bits into a symbol; a couplingelement to electromagnetically transfer of a portion of energyassociated with the symbol to a bus without significantly changing theimpedance of the bus; and a clamp to secure the memory module to asystem to which the portion of the symbol energy to be transferred. 10.The memory module of claim 9, further comprising a receiver to decodeplural bits from symbol energy received through the coupling element andto provide the decoded plural bits to the memory device for storage. 11.The memory module of claim 10, wherein the transmitter and the receiverform an interface to transfer data to and, from the memory device usingtwo or more of pulse-width modulation, phase modulation or amplitudemodulation.
 12. The memory module of claim 9, wherein the clamp securesthe coupling element in a nominal position relative to the system. 13.The memory module of claim 12, wherein the coupling element has ageometry that reduces sensitivity of the portion of symbol energytransferred to variations in the nominal position.
 14. The system ofclaim 13, wherein the coupling element has a zig-zag geometry.
 15. Anapparatus comprising: a transmitter, to encode a first plurality of bitsreceived by the apparatus into a symbol; a receiver to decode atransferred symbol received by the, apparatus into a second plurality ofbits; and a coupling element having, a geometry to enable asubstantially reliable signal transfer between the coupling element and,a bus without significantly changing the impedance of the bus; asubstrate on which the transmitter, receiver, and coupling element aremounted, the substrate including a flexible portion on which thecoupling element is mounted.
 16. The apparatus of claim 15, furthercomprising a device to provide the first plurality of bits to thetransmitter and to receive the second plurality of bits from thereceiver.
 17. The apparatus of claim 15, wherein the geometry of thecoupling element is a zig-zag geometry.
 18. The apparatus of claim 15,wherein the transmitter and the receiver support two or more modulationschemes selected from the list comprising phase modulation, pulse-widthmodulation, amplitude modulation, or rise-time modulation.
 19. Theapparatus of claim 18, wherein the selected modulation schemes aredetermined according to bit-intervals associated with each modulationscheme and a targeted symbol period.
 20. The apparatus of claim 15,wherein the coupling element has a length-induced bandwidth of at least5 Ghz.